Circuit and a method for extending the output voltage range of an integrator circuit

ABSTRACT

A circuit extends the output voltage range of an integrator circuit wherein the input signal is used to produce an output signal, and the voltage of the output signal develops monotonically within a predetermined range of possible values. The integrator circuit is driven within an integration time period such that each time the signal at its output reaches a limit of the range of values, the integrator circuit starts a subsequent integration stage of the input signal in which the output signal develops again within the above-mentioned range. This takes place by resetting the integrator circuit or by a reversal of the characteristic slope of the output signal. This is combined with storing the number of occasions on which these interventions have occurred as determined by a counter. This enables the actual voltage value of the signal resulting from the integration to be calculated by a relatively straightforward mathematical operation from the reading of the counter, and from the signal currently present at the output of the integrator at the end of the integration period.

FIELD OF THE INVENTION

[0001] The present invention relates to a circuit and to a method for extending the range of the output voltage of an integrator circuit beyond its supply voltage. More particularly, the present invention relates to a circuit of this type associated with an integrator circuit used in automotive applications, and, more specifically, in detecting knocking in internal combustion engines.

BACKGROUND OF THE INVENTION

[0002] In a system for detecting knocking in an internal combustion engine, one or more wide-band accelerometric knock sensors are provided, and advantageously are disposed on the engine block in the vicinity of the cylinders. These sensors register variations in pressure on the cylinder walls and translate them into electrical signals which are processed in a control unit to distinguish the pressure contributions due to knocking from those relating to operation with correct combustion.

[0003] During this processing, the electrical signal coming from the sensor is amplified and filtered, and after being rectified, is sent to an integration stage which outputs a voltage signal. This voltage signal is proportional to the energy of the initial electrical signal, is within the filtering band, and is proportional to the integration period.

[0004] At the end of the integration period, the value of the voltage signal reached by an integrator circuit of the integration stage is stored, for example, in a sample/hold circuit and made available as an output to further processing stages. These further processing stages are arranged to identify the occurrence of knocking from the value of this signal and to provide feedback control to a system controlling ignition in the engine.

[0005] It can easily be understood that the value of the voltage signal output by the integrator circuit may reach high levels if the integration time is long. Conventional integrator circuits formed with operational amplifiers and capacitive feedback components have a maximum limit for their output voltage, which may increase or decrease monotonically within the integration time period. This limit cannot be passed and is determined by the supply voltage supplied to the circuit, or by the supply voltages if there are two, that is, one positive and one negative.

[0006] When an operational amplifier is required to have an output voltage close to or greater than this limit, it ceases to operate linearly and reaches a saturation condition in which the voltage no longer increases (decreases) as the integration time passes. Instead, the voltage adopts a maximum (minimum) limit value which is substantially constant and is within the limits imposed by the supply voltage.

[0007] The approaches according to the prior art, which are referred to in the technical literature as “rail-to-rail” circuits, do not provide for these limits to be exceeded, but only to be approached as closely as possible.

SUMMARY OF THE INVENTION

[0008] An object of the present invention is to provide a system which enables the range of the output voltage of an integrator circuit to be artificially extended beyond the limits imposed by the supply voltage.

[0009] According to the present invention, this object is achieved by a circuit having the characteristics recited in claim 1. A further subject of the invention is a method having the characteristics recited in claim 12.

[0010] In summary, the present invention is based on the principle of monitoring the development of the voltage signal generated by an integrator circuit according to the prior art and resetting the circuit (or, in an alternative embodiment, reversing the characteristic slope of the output signal) each time its output voltage reaches a predetermined limit close to the saturation condition.

[0011] This is combined with the step of memorizing the number of occasions on which these interventions have occurred by using a counter which is connected to the integrator circuit, and which is incremented each time the integrator is reset (or the slope of the output signal is reversed).

[0012] At the end of the predetermined integration period, the content of the counter will thus indicate how many times the voltage signal generated by the integrator has covered the entire range naturally available during its increasing or decreasing development. This will enable the actual voltage value of the signal resulting from the integration to be calculated by a simple mathematical operation from the reading of the counter and from the signal currently present at the output of the integrator, as will be described further in the following examples.

[0013] The embodiment according to the present invention thus enables a substantially unlimited, although fictitious, output voltage range to be provided in an integrator circuit. In a circuit of the type used, for example, in control systems for detecting the degree of knocking in internal combustion engines, this enables simpler control systems to be produced. These control systems are advantageously produced which operate from a single voltage supply (for example, 5 V) both for the active elements of the integrator circuit and for the logic circuits, and any micro-controllers present in the engine electronic control unit.

[0014] A further advantage is that the integration period can be extended at will and more efficient engine control algorithms can be established. The greater efficiency achieved enables an engine of the same type to have lower consumption and greater power than with current approaches.

BRIEF DESCRIPTION OF THE DRAWINGS

[0015] Further characteristics and advantages of the invention will be explained in greater detail in the following detailed description of different embodiments thereof, given by way of non-limiting examples, with reference to the appended drawings, in which:

[0016]FIG. 1 is a block diagram of a system for detecting knocking in an internal combustion engine according to the present;

[0017]FIG. 2 is a circuit diagram of a first embodiment of an integration circuit stage comprising a circuit according to the present invention,

[0018]FIG. 3 is a series of graphs indicative of the quantities representative of the operation of the integration circuit stage illustrated in FIG. 2;

[0019]FIG. 4 is a circuit diagram of a second embodiment of an integration circuit stage comprising a circuit according to the present invention; and

[0020]FIG. 5 is a series of graphs indicative of the quantities representative of the operation of the integration circuit stage illustrated in FIG. 4.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0021] In a system for detecting knocking in an internal combustion engine, a wide-band accelerometric knock sensor 10 is disposed on the engine block in the vicinity of the cylinders. The sensor 10 registers variations in pressure on the walls of the cylinders and translates them into an electrical voltage signal, which is indicated as V_(k) in the circuit diagram of FIG. 1 and is proportional to the acoustic energy detected.

[0022] An amplifier block 12 is coupled to the sensor 10 and receives and amplifies the signal V_(k).A band-pass filter 14 previously tuned to the characteristic knock frequency of the engine in question is downstream of the amplifier block 12. The amplified and filtered voltage signal V_(F) is a signal proportional to the amplitude of the knocking alone. A rectifier stage 16 is downstream of the filter 14 and has an output voltage signal V_(r) substantially corresponding to the envelope of the signal V_(F).

[0023] An integration stage 18 is coupled to the output of the rectifier stage 16 by a first input (or signal input) and has its output connected to a first input of a comparator circuit 20. The output of the comparator circuit 20 corresponds to the output of the knocking detection system as a whole.

[0024] The integration stage 18 comprises a conventional operational-amplifier integrator circuit 22, a circuit for extending the output voltage range of the integrator circuit, and a sample/hold circuit 24. The sample/hold circuit 24 is for the temporary storage of the voltage value reached at the output of the integrator circuit 22 at the end of the integration period. This value is proportional to the knock energy.

[0025] When the system is in operation, the signal V_(r) output by the rectifier stage 16 is integrated within a predetermined period of time T₁ to derive an output signal V_(out). The integration period Ti is determined by a control logic signal GATE supplied to a second input (or control input) of the integration stage 18 so as to correspond to the period of time in which the intensity of knocking is greater than the background noise. The selection of the duration of this period of time determines the efficiency of the knocking detection.

[0026] The comparator circuit 20 has a second input which receives a reference signal V_(th) indicative of a knock intensity threshold, and is arranged to emit an output signal V_(c) indicative of the occurrence of knocking. As is well known to one skilled in the art, this signal may be made available to a circuit controlling ignition advance in a conventional closed-loop system.

[0027] A first embodiment of an integration stage 18 according to the invention is described in detail with reference to FIG. 2. The integrator circuit 22 receives the rectified voltage signal V_(r) by a first input (or signal input) of the integration stage 18, and has its output coupled to the sample/hold circuit 24. The control input of the sample/hold circuit 24 receives the control logic signal GATE from the second input (or control input) of the integration stage.

[0028] The output of the sample/hold circuit 24, at which the output signal V_(o) is produced by the integrator circuit 22, is connected to the inverting input of a threshold comparator 30. The non-inverting input of the threshold comparator 30 receives a reference voltage V_(sup). The function of the comparator 30 is to detect when the voltage of the signal V_(o) exceeds the value V_(sup).

[0029] The output of the threshold comparator 30 is connected to the control input TR1 _(i) of a first monostable multivibrator 32, the transition of which from the stable state to the quasi-stable state is induced by the trailing edge of the signal output by the comparator. The multivibrator 32 in turn is coupled to a resetting input of the integrator circuit 22 by an AND logic gate 34.

[0030] The control logic signal GATE is present at the control input TR2 _(i) of a second monostable multivibrator 36, the transition of which from the stable state to the quasi-stable state is induced by the leading edge of the signal GATE. The output TR2 _(o) of the multivibrator 36 is also coupled to the resetting input of the integrator circuit 22 by the AND logic gate 34.

[0031] A counter 38 is coupled to the output TR2 _(o) of the second multivibrator 36 by its own resetting input CL, and has its own drive input CK coupled directly to the output of the comparator 30. The increment of the counter is induced by the trailing edge of the signal present at the drive input. A plurality of output terminals b0, b1, . . . , bn, and OV is provided for presenting the content of the counter and for indicating a possible overflow condition thereof, respectively. The monostable multivibrators 32, 36 and the counter 38 are together indicated as the control circuit 40 of the integrator circuit 22.

[0032] The operation of the integration stage 18 as a whole will now be described in detail with reference to FIG. 3. At the beginning of an integration period T₁ determined by the signal GATE at a high logic level, both the integrator circuit 22 and the sample/hold circuit 24 are reset. The resetting of the integrator circuit 22 is controlled by a signal RESET which has a pulse (logic zero) of predetermined time duration produced by the multivibrator 36, and which is excited by the active edge of the signal GATE.

[0033] The sample/hold circuit 24 is operated in its sampling stage so as to transmit to its output the signal currently present at the output of the integrator V_(o). At the same time, the content of the counter 38 is cleared. The curves of the signals GATE and RESET against time are shown in the third and fifth graphs of FIG. 3, respectively.

[0034] In the example shown, the signal V_(r) input to the integrator circuit 22 is a rectified sinusoidal signal. The integrator circuit 22 generates a substantially ramp-like output signal V_(o). When this signal has reached the reference voltage V_(sup) it causes the output signal COMP of the threshold comparator 30 to switch (see the fourth graph of FIG. 3) from a first logic level (high level) to a second logic level (low level).

[0035] The trailing edge of the signal COMP increases the content of the counter, and at the same time, brings about the transition of the multivibrator 32 to the quasi-stable state so that the multivibrator 32 sends a resetting pulse to the integrator circuit 22. When resetting has taken place, the signal V_(o) can start to increase again in dependence on the signal V_(r), and obviously the output signal COMP of the threshold comparator 30 switches back to the first logic level.

[0036] The process described may be repeated any number of times, basically in dependence on the preselected duration of the integration time period T_(i). The limit of the operation of the control circuit 40 is imposed exclusively by the capacity of the counter 38 used. An overflow condition of the counter may be indicated by a high level signal at the output terminal OV.

[0037] At the end of the integration period T_(i), the control signal GATE switches to a low logic level and the sample/hold circuit 24 stores the voltage value V_(o) reached at that moment at the output of the integrator circuit 22.

[0038] The actual value of the voltage of the output signal V_(out) of the integration stage 18 can thus be derived mathematically from the voltage value V_(o) reached at the output of the integrator circuit at the end of the integration period (final value) and from the content of the counter. That is, from the number N of times (encoded in binary form by the bits b0, b1, . . . , bn) the signal V_(o) has covered the entire range available at the output of the integrator circuit during its increasing development. This value is given by the expression:

V _(out) =N×V _(sup) +V _(o)

[0039] The variable V_(sup) is the value of the reference voltage of the threshold comparator corresponding to the upper value of the output voltage range of the integrator circuit.

[0040] This operation can easily be performed by a conventional processing unit (not shown) arranged to convert the analog signal V_(o) into a digital signal, and to perform the programmed arithmetic calculation.

[0041] In an alternative embodiment, the control circuit 40 may be implemented as a single digital circuit, for example, a finite state machine. This finite state machine is arranged to receive the input signals GATE and COMP, and advantageously generates the resetting signals for the integrator circuit 22, the control signals for the sample/hold circuit 24, and the output signal of the counter 38 while maintaining substantially the same timings as described above.

[0042] With the circuit arrangement described by way of example, the output voltage range of the integration stage 18 is thus extended artificially by resetting the integrator circuit 22 each time its output voltage reaches a predetermined limit close to the saturation condition, and by counting the number of resettings.

[0043] Alternatively, an approach is provided in which the range is extended by varying the gain of the integrator circuit so as to reverse the characteristic slope of its output signal each time its output voltage reaches a predetermined upper or lower limit close to the saturation condition, and by counting how many times this reversal operation has taken place.

[0044] A second embodiment which can bring about this behavior is described in detail with reference to FIG. 4. Elements identical or functionally equivalent to those illustrated in FIG. 2 have been indicated by the same references already used in the description of the previous embodiment.

[0045] The integrator circuit 22 is coupled to an amplifier block 50 with unitary gain. The input of the amplifier block 50 receives the rectified voltage signal Vr that is also input to the integration stage. The integrator circuit 22 is also coupled to the sample/hold circuit 24 which is downstream therefrom, and the control input of which receives the control logic signal GATE from the second input of the integration stage 18.

[0046] The task of the amplifier block 50 is simply to transfer the signal V_(r) input to the integrator circuit 22 in a direct or inverted manner, advantageously establishing a gain of +1 or −1, as required. The block 50 may be formed as a set of two amplifiers with unitary gain of the inverting type and of the non-inverting type, respectively, which can be selected by associated switches. Alternatively, the block 50 may be incorporated in the integrator circuit 22 if it is of the type with switched capacitors so that the selection of a positive or negative gain takes place by suitable driving of the switches provided.

[0047] The output of the sample/hold circuit 24 provides the signal V_(o) which is applied to the non-inverting input of a first threshold comparator 52 which has an inverting input receiving a first reference voltage V_(sup), and to the inverting input of a second threshold comparator 54 which has a non-inverting input receiving a second reference voltage V_(inf). The function of the comparators 52 and 54 is to detect when the voltage of the signal V_(o) exceeds the value V_(sup) or falls below the value V_(inf), respectively.

[0048] The outputs of the first and second threshold comparators 52, 54 are connected, respectively, to the resetting input R and to the control input S of a bistable multivibrator or RS flip-flop 56 which in turn is connected to a control input of the amplifier block 50.

[0049] The control logic signal GATE establishes at the control input TR₁ of a monostable multivibrator 36, the transition of which from the stable state to the quasi-stable state is induced by the leading edge of the signal GATE. The output TR_(o) of the multivibrator 36 is coupled to the resetting input of the integrator circuit 22.

[0050] A counter 38 is coupled to the output TR_(o) of the multivibrator 36 by its own resetting input CL and has its own drive input CK coupled to the outputs of the comparators 52, 54 by an OR logic gate 58. The increment of the counter is induced by the leading edge of the signal present at the drive input. A plurality of output terminals b0, b1, . . . , bn, and OV is provided for presenting the content of the counter and for indicating a possible overflow condition thereof, respectively.

[0051] The monostable multivibrator 36, the flip-flop 56, and the counter 38 are together indicated as the control circuit 40 of the integrator circuit 22. The operation of the integration stage 18 as a whole, according to this embodiment will now be described in detail with reference to FIG. 5.

[0052] At the beginning of an integration period Ti, determined by the signal GATE at a high logic level, both the integrator circuit 22 and the sample/hold circuit 24 are reset. The resetting of the integrator circuit is controlled by the signal RESET which has a pulse (logic one) of predetermined time duration. This pulse is produced by the multivibrator 36 excited by the active edge of the signal GATE. The sample/hold circuit 24 is operated in its sampling stage as in the previous embodiment. At the same time, the content of the counter 38 is cleared. The curves of the signals GATE and RESET against time are shown in the third and seventh graphs of FIG. 5, respectively.

[0053] As a result of the resetting of the integrator circuit, the output signal V_(o) initially has a voltage lower than both of the reference voltages V_(inf) and V_(sup), so that the signal C2 output to the threshold comparator 54 is at a high logic level and brings about, by the control input S, the transition of the flip-flop 56 to a first state. In this state, the flip-flop emits a control logic signal SGAIN to the amplifier block 50 such as initially to establish a positive gain value (for example, SGAIN=1).

[0054] It will be clear to one skilled in the art, however, that this transition of the signal C2 is not noticed at the drive input CK of the counter 38 since the latter simultaneously receives a resetting signal at the input CL.

[0055] In the embodiment shown, the signal V_(r) input to the integrator circuit 22 is again a rectified sinusoidal signal so that the integrator circuit generates a substantially ramp-like output signal V_(o). When this signal has reached the first reference voltage V_(sup), it causes the output signal Cl of the threshold comparator 52 to switch (see the fourth graph of FIG. 5) from a first logic level (low level) to a second logic level (high level).

[0056] The leading edge of the signal Cl increases the content of the counter 38, and at the same time, brings about by the resetting input R the transition of the flip-flop 56 to a second state in which the flip-flop consequently emits a control logic signal SGAIN to the amplifier block 50 such as to establish a negative gain value (for example, SGAIN=0).

[0057] When the gain of the amplifier block has been changed, the signal V_(o) develops with a decreasing amplitude, again in dependence on the signal V_(r). The output signal Cl of the threshold comparator 52 switches back to the first logic level. When the signal V_(o) reaches the second reference voltage V_(inf), it causes the output signal C2 of the threshold comparator 54 to switch (see the fifth graph of FIG. 5) from a first logic level (low level) to a second logic level (high level).

[0058] The leading edge of the signal C2 increases the content of the counter 38 again, and at the same time brings about by the control input S the transition of the flip-flop 56 to the first state. This causes the emission of a control signal SGAIN to the amplifier block 50 such as to re-establish a positive gain value (for example, SGAIN=1 again).

[0059] The process described above may be repeated any number of times in dependence on the preselected duration of the integration period T_(i). As in the previous embodiment, the limit of the operation of the control circuit 40 is imposed exclusively by the capacity of the counter 38 used, and any overflow condition is indicated by a high level signal at the output terminal OV. At the end of the integration period T_(i), the control signal GATE switches to a low logic level and the sample/hold circuit 24 stores the voltage value V_(o) reached at that moment at the output of the integrator circuit 22.

[0060] The actual value of the voltage of the output signal V_(out) of the integration stage 18 can thus be derived mathematically from the voltage value V_(o) reached at the output of the integrator circuit at the end of the integration period (final value) and from the content of the counter. That is, from the number N of times (encoded in binary form by the bits B0, b1, . . . , bn) the signal V_(o) has covered the entire range available at the output of the integrator circuit during its increasing and decreasing development. This value is given by the expression:

V _(out) =N×(V _(sup) −V _(inf))+(V _(sup) −V _(o))×SGAIN+ (V _(o) −V _(inf))×(1−SGAIN)+V _(inf)

[0061] The variable V_(sup) is the value of the first reference voltage of the threshold comparator 52 corresponding to the upper value of the output voltage range of the integrator circuit. The variable V_(inf) is the value of the second reference voltage of the threshold comparator 54 corresponding to the lower value of the output voltage range of the integrator circuit. The variable SGAIN is the logic value adopted by the control signal of the amplifier at the end of the integration period. That is, 1 if the gain of the amplifier is positive (+1) and 0 if it is negative (−1).

[0062] In the above expression, the last addition takes account of the fact that, initially, when the integrator circuit 22 is reset, its output voltage V₀ adopts a substantially zero starting value which is generally different from the value of the second reference voltage value V_(inf). As already described with reference to the previous embodiment, this operation can easily be performed by a conventional processing unit (not shown).

[0063] In an alternative embodiment, the control circuit 40 may be implemented as a single digital circuit, for example, a finite state machine arranged to receive the input signals GATE, C1 and C2. The finite state machine advantageously generates the resetting signal for the integrated circuit 22, the control signal for the sample/hold circuit 24, the control signal for the amplifier block 50, and the output signal of the counter 38 while maintaining substantially the same timings as described above.

[0064] It is clear from the examples described that, with the use of a circuit according to the invention, it is possible to advantageously use a low supply voltage, for example 5 V, for the active elements of the integrator circuit. By using “rail-to-rail” integrator circuits, it is possible to set values of about 0.5 V and 4.5 V for the reference voltages V_(inf) and V_(sup), respectively, but without limiting the range of the voltage V_(out) which can be reached by the integration stage as a whole to 4 V.

[0065] Naturally, the principle of the invention remains the same, the forms of embodiment and details of implementation may be varied widely with respect to those described and illustrated purely by way of non-limiting examples, without thereby departing from the scope of protection of the present invention. In particular, although the examples relate to embodiments in which the output voltage of the integrator circuit adopts exclusively positive values, one skilled in the art will have no problem in appreciating, in the light of the foregoing description, that these embodiments may be extended to a situation in which the integrator circuit has a symmetrical dual supply. 

That which is claimed is:
 1. A circuit for extending the output voltage range of an integrator circuit wherein the input signal is such as to produce an output signal the voltage of which develops monotonically within a predetermined range of possible values, said circuit comprising: control circuit means for controlling the integrator circuit in a manner such that, within an integration time period, each time the output signal of the integrator circuit reaches a limit of the range of values, the integration circuit starts a subsequent integration stage of the input signal in which the signal output by the integrator develops again within the same range; and counting means associated with the control means for counting and outputting the number of times the output signal of the integrator circuit has reached a limit value, the number being correlated with the number of times the output signal has covered the said range of values, the actual voltage value reached at the end of the integration period being calculable from a final voltage value of the output signal of the integrator circuit reached at the end of the integration period, and from the number of times the signal has reached a limit value.
 2. A circuit according to claim 1 , wherein the control circuit means comprise: comparator means for comparing the voltage of the output signal of the integrator circuit with at least one reference voltage corresponding to a limit of the range of values, the means being able to emit at least one respective control signal as a result of the reaching of said limit during the development of the output signal; and drive means for driving the integrator circuit, the drive means being associated with the comparator means and being arranged to cause a subsequent integration stage of the input signal to start in dependence on the control signals received from the comparator means.
 3. A circuit according to claim 2 , wherein the comparator means comprise a threshold comparator circuit to the non-inverting input of which a reference voltage is applied and to the inverting input of which the output signal of the integrator circuit is applied, and the drive means comprise a monostable multivibrator circuit which is arranged to receive, at a control input, a control signal emitted by the threshold comparator circuit, and which can emit, towards the integrator circuit, a resetting pulse of predetermined duration each time the comparator circuit detects that the output signal has reached a limit value.
 4. A circuit according to claim 3 , wherein the counting means can be coupled at their output to a processing unit which can read its content and which can receive a datum relating to the final voltage value of the output signal of the integrator circuit, the processing unit being arranged to calculate the actual voltage value reached at the end of the integration period in accordance with the equation: V _(out) =N×V _(sup) +V _(o) in which: V_(sup) is the value of the reference voltage of the threshold comparator.
 5. A circuit according to claim 2 , wherein the comparator means comprise: a first threshold comparator circuit to the inverting input of which a first reference voltage corresponding to an upper limit of the range of possible values for the output voltage signal of the integrator circuit is applied, and to the non-inverting input of which the voltage of the output signal of the integrator circuit is applied, and a second threshold comparator circuit to the non-inverting input of which a second reference voltage corresponding to a lower limit of the range of possible values for the output signal of the integrator circuit is applied, and to the inverting input of which the voltage of the output signal of the integrator circuit is applied.
 6. A circuit according to claim 5 , comprising control circuit means for controlling the gain of the integrator circuit, the control circuit means being associated with the integrator circuit and being arranged to invert its gain in dependence on a control input signal.
 7. A circuit according to claim 6 , wherein the drive means comprise a bistable multivibrator circuit which is arranged to receive, at its control input and at its resetting input, control signals emitted by the first and by the second threshold comparator circuits, respectively, and which can emit towards the gain control means a respective control signal such as to invert the gain each time a comparator circuit detects that the voltage of the output signal has reached a limit value, so as to reverse the trend of the output signal of the integrator circuit.
 8. A circuit according to claim 6 , wherein the gain control means of the integrator circuit comprise a pair of amplifier circuits with unitary gain of the inverting and non-inverting type, respectively.
 9. A circuit according to claim 7 or claim 8 , wherein the counting means can be coupled at their output to a processing unit which can read its content and which can receive a datum relating to the final value of the output voltage signal of the integrator circuit, the processing unit being arranged to calculate the actual voltage value reached at the end of the integration period in accordance with the equation: V _(out) =N×(V _(sup) −V _(inf))+(V _(sup) −V _(o))×SGAIN+(V _(o) −V _(inf))×(1−SGAIN)+V _(inf) in which: V_(sup) is the value of the first reference voltage of the respective first threshold comparator, V_(inf) is the value of the second reference voltage of the respective second threshold comparator, and SGAIN is the logic value adopted by the control signal of the gain control means of the integrator circuit at the end of the integration period.
 10. A circuit according to claim 1 , further comprising resetting means connected to a resetting input of the integrator circuit and arranged to receive, at a control input, a signal for controlling the integration operation and to emit, towards the integrator circuit, a resetting signal.
 11. A circuit according to claim 10 , wherein the resetting means comprise a monostable multivibrator circuit which can emit a resetting pulse of predetermined duration towards the integrator circuit at the beginning of an integration period.
 12. A method of extending the range of the output voltage of an integrator circuit wherein the input signal is such as to produce an output signal the voltage of which develops monotonically within a predetermined range of possible values, the method comprising the steps of: comparing the voltage of the output signal of the integrator circuit with at least one reference voltage corresponding to a limit of the range of values, driving the integrator circuit, after the limit has been reached during the development of the output signal during an integration period, in a manner such that said circuit starts a subsequent integration stage of the input signal in which the output signal develops again within the same range, counting the number of times the output signal of the integrator circuit has reached a limit value so that a subsequent integration stage has been started, the number being correlated with the number of times the output signal has covered the range of values, and calculating the actual voltage value reached at the end of the integration period from a final value of the voltage of the output signal of the integrator circuit and from the number of times the signal has reached a limit value.
 13. A method according to claim 12 , wherein the step of driving the integrator circuit includes resetting of the circuit each time its output signal has reached a limit value of the range of possible values.
 14. A method according to claim 13 , wherein the actual voltage value reached at the end of the integration period is calculated in accordance with the equation: V _(out) =N×V _(sup) +V _(c) in which: V_(sup) is the value of the reference voltage.
 15. A method according to claim 12 , wherein the voltage of the output signal of the integrator circuit is compared with a pair of reference voltages, that is, a first reference voltage corresponding to an upper limit of the range of values and a second reference voltage corresponding to a lower limit of the range of values, respectively.
 16. A method according to claim 15 , wherein the step of driving the integrator circuit includes the inversion of the gain of the circuit each time the voltage of the output signal has reached one of the upper and lower limit values, so as to reverse the trend of the signal.
 17. A method according to claim 16 , wherein the actual voltage value reached at the end of the integration period is calculated in accordance with the equation: V _(out) =N×(V _(sup) −V _(inf))+(V _(sup) −V _(o))×SGAIN+(V _(o) −V _(inf))×(1−SGAIN)+V _(inf) in which: V_(sup) is the value of the first reference voltage, V_(inf) is the value of the second reference voltage, and SGAIN is a logic value determined in dependence on the gain of the integrator circuit at the end of the integration period. 